By raising the switching frequency used in AC-DC converters which obtain a desired DC power from a commercial power supply, it is possible to reduce the sizes of the transformers, inductors, and other components used; but as the switching frequency is raised, increased switching losses and other circuit losses pose problems, and various circuit-related innovations are being made to improve efficiency.
Current resonance (multi-resonance) type AC-DC converters, such as that described in Patent Document 1, are known as insulation type AC-DC converter technology of the prior art which can suppress switching losses even at higher frequencies. FIG. 10 is a block diagram showing the electrical configuration of such a multi-resonance type AC-DC converter 1. This converter 1 substantially comprises a diode bridge db, smoothing capacitor c1, and DC-DC converter 2. The sinusoidal alternating current voltage Vac from the commercial power supply 3 is input via a current fuse f to the diode bridge db and smoothing capacitor c1, and a rectified and smoothed DC voltage is output as the power supply voltage of the DC-DC converter 2.
In the DC-DC converter 2, the power supply voltage is applied to the two-stage series switching elements q1 and q2; connected in parallel with one of the switching elements q2 are a series resonance circuit, comprising a choke coil 11, a primary winding t1 of an insulation transformer t, and capacitor c2, as well as a capacitor c3. The secondary winding t2 of the insulation transformer t are connected, via the diodes d1 and d2 respectively, to the high-side terminal of a smoothing capacitor c4, and a center tap is connected to the low-side terminal of the smoothing capacitor c4. By means of this DC-DC converter 2, the desired DC voltage, rectified and smoothed, is supplied to the DC load 4.
FIG. 11 shows waveforms at various portions, for use in explaining the operation of the above circuit of the prior art. Vg1 and Vg2 are gate signals applied by the control circuit 5 to the MOSFETs q1, q2 which are switching elements. In response to these gate signals Vg1, Vg2, the switching elements q1, q2 are turned on and off in alternation, and the drain-source voltages and drain currents thereof assume the waveforms Vq1, Iq1 and Vq2, Iq2, respectively. Vc2 is the voltage applied to the capacitor c2; by setting the series circuit to appropriate LC series resonance conditions for the switching frequency, a substantially sinusoidal current resonance state results.
Id1 and Id2 are the current waveforms of the diodes d1, d2 on the secondary side of the insulation transformer t; there exist conducting and non-conducting intervals with the timing shown, due to the difference between the voltage induced on the secondary side of the insulation transformer t and the DC voltage of the smoothing capacitor c4. During an interval in which a diode d1 or d2 is conducting, the secondary side of the insulation transformer t is in a short-circuited state via the diode d1 or d2. If for simplicity the insulation transformer t is assumed to be a non-gap transformer (with close coupling between the primary winding t1 and the secondary winding t2), then the primary-side excited inductance of the insulation transformer t is also substantially short-circuited, so that the inductor 11 and capacitor c3 undergo series resonance. On the other hand, during intervals in which both diodes d1 and d2 are non-conducting the secondary side of the insulation transformer t is in the open-circuit state, and the capacitor c3 and combined value of the inductor 11 and the transformer excited inductance (10) undergo series resonance.
Hence the resonance frequency f1 of the circuit in the interval W1 in which diode d1 or d2 is conducting is ½π(11·c3)1/2, and the resonance frequency f2 in the non-conducting interval W2 is ½π((11+10)·c3)1/2. Hence the resonance frequency f2 is lower than the resonance frequency f1. In FIG. 11, td is the dead-off time, and T is one period.
From these operation waveforms, when the switching element q1 or q2 is turned on, the switching current is a somewhat negative current (flowing in the internal diode of a MOSFET), so that zero-current switching (ZCS) operation is possible, and switching losses are extremely low. And, when the switching element q1 or q2 is turned off, during the dead-off interval the capacitor c3 connected in parallel with the switching element q2 absorbs resonance energy of the inductor 11, and the applied voltage rises with a gentle gradient, so that zero-voltage switching (ZVS) operation by soft switching is possible, and switching losses are extremely low.
In such a multi-resonance type AC-DC converter 1, there is suppression of the increase in switching losses which is a concern when the switching frequency is raised, so that such a design is well-suited to miniaturization. However, in this converter 1 the control circuit 5 monitors the output voltage via a feedback circuit 6, and when load fluctuation occurs, the switching frequency is changed to perform fluctuation compensation in order to hold the output constant while maintaining multi-resonance waveforms. As a result, when an attempt is made to compensate the output in response to large-amplitude load fluctuations, fluctuations in the voltage of the commercial power supply, or other broad-range fluctuations, it becomes extremely difficult to maintain multi-resonance waveforms, and ultimately there is the problem that device selection and measures to address heat dissipation are indispensable in order to address deviation from the multi-resonant state. There is also a problem of imparting harmonic distortion to input current from the commercial power supply. Regulations regarding harmonic distortion are particularly strict in illumination applications.
FIG. 12 shows a standard AC-DC converter 11 with improved power factor designed to resolve such problems. This converter 11 substantially adopts the configuration of the above-described AC-DC converter 1, but with a filter circuit comprising an inductor 12 and capacitor c5 inserted on the input side of the diode bridge db, and with the ripple current full-rectified by the diode bridge db stepped up as-is by the boosting chopper circuit 12. And, the DC voltage resulting from smoothing by the smoothing capacitor becomes the power supply voltage of the step-down DC-DC converter 2.
The boosting chopper circuit 12 applies the ripple current output voltage from the diode bridge db to the series circuit of the choke coil 13, switching element (MOSFET) q3, and the source resistance thereof r. Through switching of the switching element q3 by the control circuit 13, a stepped-up voltage is extracted from the connection point of the choke coil 13 and the switching element q3, and is applied via the diode d3 to the smoothing capacitor c1. The control circuit captures input voltage signals, output voltage feedback signals, switching current signals, and synchronization signals (signals from auxiliary winding of the choke coil 13), controls the chopper switching element q3 such that the switching current value coincides with a reference value obtained from the product of the input voltage signal and the output voltage feedback signal, and through the effect of the filter circuit provided at the commercial power supply 3 and comprising the choke coil 12 and the capacitor c5, obtains a sinusoidal input current.
Through this configuration, an AC-DC converter with a higher input voltage to the DC-DC converter 2, with harmonic distortion suppressed in the input alternating current voltage Vac from the commercial power supply 3 and with a high power factor, can be realized. However, there are the problems of increases losses in the circuit as a whole due to cascade connection of two converters (12, 2), and of increases in cost and diminished advantages of miniaturization accompanying an increase in the number of components. By providing a boosting chopper circuit 12 in the first stage of the DC-DC converter 2, the input voltage is stabilized and there is no need for compensation for fluctuation in the voltage of the commercial power supply 3, and to this extent control by the control circuit 5 is facilitated.
FIG. 13 shows an AC-DC converter 21 with the cascade configuration of the two converters in the above configuration interchanged. This technology of the prior art is disclosed in Patent Document 2. In this converter 21, the output of the first-stage converter 22 is input to the converter of the second stage (boosting chopper circuit) 23 without smoothing. That is, in this AC-DC converter 21, the input alternating current voltage Vac from the commercial power supply 3 is rectified by the diode bridge db1, the resulting ripple current is input to the first-stage converter 22, conversion to a high-frequency AC voltage is performed by full-bridge inverter switching by means of the switching elements q11 to q14, output with the voltage changed by the insulation transformer t is obtained, and this output is again rectified by the diode bridge db2, and after passing through the second-stage converter 23 a DC output is obtained. The control circuit 24 of the converter executes control such that the AC input current iac is sinusoidal corresponding to the input alternating current voltage Vac, and such that the DC output voltage VA is constant.
The converter 23 applies the high-frequency ripple output voltage from the diode bridge db2 to the series circuit of the chopper coil 14 and switching element q3, and through switching of the switching element q3 by the control circuit 24, a stepped-up voltage is obtained from the connection point of the choke coil 14 and the switching element q3; this voltage is applied to the DC load 4 from the capacitor c6 via the diode d3.
Features of the AC-DC converter 21 shown in FIG. 13 are, together with measures addressing harmonic distortion, the ability to eliminate the high-voltage, high-capacitance smoothing capacitor c1 on the input side; the elimination of the need for measures to deal with inrush current at power-on as a result; and, the fact that the chopper coil 14 and capacitor c6 are used in common as the smoothing filter of the first-stage converter 22 and the smoothing filter of the second-stage converter 23.
However, there are the problems that no innovations are made to reduce losses in the first-stage full-bridge inverter (22), and that the overall efficiency is reduced by the cascade connection with the second-stage converter 23. Further, because the input current to the second-stage converter 23 is sinusoidal, the control circuit 24 must monitor the primary-side AC input current iac and input alternating current voltage Vac as well as the secondary-side DC output voltage VA of the insulation transformer t, so there is a need for a current transformer, voltage transformer, or other insulating means, and problems with respect to costs and shape occur.    Patent Document 1: Japanese Patent No. 3371595    Patent Document 2: Japanese Patent No. 2514885